Modern power converter systems often use various switched-mode power supplies (SMPS) to convert a DC voltage (or also an AC voltage, if the coupling includes suitable rectifying means) to a more suitable voltage level and to stabilize the voltage and current fed to the load in conditions where the supply power and the electrical characteristics of the load may vary. In many different applications it is necessary to restrict the output current and voltage of the power supply below a certain maximum value so as not to harm the load and the power supply itself. In addition, it is often advantageous that the output voltage and current can be simultaneously controlled in such a way that there is a relationship between their values. An example of such an application is a charging device for charging batteries. When a discharged battery is coupled to the charging device, its terminal voltage is relatively low at first. During charging, the terminal voltage increases towards a maximum value which depends on the materials and construction of the battery. The temperature of the charging device increases with the terminal voltage of the battery, and the charging current must be limited so as to avoid disadvantageous phenomena, such as excessive warming. Towards the end of the charging process, the current reception capability of the battery fails, whereupon it is advantageous to reduce the output current of the charging device and to limit its output voltage below a certain maximum value so as to avoid losses and other disadvantageous phenomena.
FIG. 1 shows, by way of example, the limit values of the output characteristics of a battery charging device, represented by the area confined by the boundary lines in the figure. The device in question is used to charge a battery, the terminal voltage of which is 10 V when fully charged. When a discharged battery is coupled to the charger, its terminal voltage rises in a few seconds to a certain minimum value, which in this case is about 5 V. The figure shows that the accurately defined current-voltage region, or the narrow "channel" between the boundary lines, starts at five volts, where the output current must be at least 0.76 A and no more than 0.84 A. Below this voltage level the output characteristics of the charger are of little importance as long the charger does not produce a current peak of more than 0.84 A upon switch-on.
During the charging process, the output current of the charger described must be kept substantially constant at about 0.8 amps. The boundary lines confine a tolerance region which represents the allowed .+-.5% variation range for the current value. It is not practical to specify the current value more accurately than this, because the electrical characteristics of components always vary within certain tolerances. The terminal voltage of the battery charged increases until it begins to near the nominal maximum value of 10.0 V. At this point, the voltage limiter of the charger starts limiting its output voltage to prevent it from exceeding the upper limit value. The voltage limiter is designed for 10.0 volts, but there is a .+-.5% variation range about the nominal value just as in the case of current limiting. When the charging process ends, the output current of the charger is substantially zero and its output voltage at least 9.5 V and not more than 10.5 V. After this, it is known to continue charging using a so-called trickle charge function, where a switch on the secondary side, i.e. on the load side of the charger switches the charging repeatedly on and off. Also in this case the limiter functions of the charger must keep the output current and voltage within allowed limits.
A switched-mode power supply which can be used to realize the charging function described above typically comprises a transformer which divides the power supply into a primary part and a secondary part. Input voltage is connected to the input terminals in the primary. Typically the power supply includes a switching element, advantageously a MOSFET or bipolar transistor which chops the input voltage into pulses that supply current to the primary winding of the transformer. Variation of the primary voltage and current stores magnetic energy in the magnetic field of the transformer. With suitable polarity in the primary and secondary windings of the transformer and using rectifier diodes the stored energy is transferred to the secondary winding and therefrom to the secondary part of the switched-mode power supply, where it produces an output voltage at the power supply output terminals. There are several known circuit topologies, or ways to arrange the components in the power supply apparatus described in relation to each other in order to achieve the desired operation. The most popular of these are the buck, boost and flyback topologies.
The traditional approach in implementing stabilization of output characteristics in a switched-mode power supply like the one described here has been to measure the output current and voltage in the secondary and take the measuring data to the circuit element in the primary that determines the duty cycle, or the ratio of the ON and OFF times of the switching transistor used to chop the primary voltage. The longer the ON time in proportion to the OFF time, the greater the amount of energy stored in the transformer's magnetic field dining an ON-OFF cycle, and the greater the amount of energy transferred through the secondary winding to the secondary of the circuit and further to the load. It depends on the construction of the secondary part and on the electrical characteristics of the load, whether it is the output voltage, the output current or both that increases. Correspondingly, decreasing the pulse ratio, or the proportion of the ON time, reduces the amount of energy transferred and hence the output voltage or the output current or both.
In a switched-mode power supply like the one described the power and voltages may be relatively high. As regards electrical safety, it is often preferable that there is no galvanic contact between the primary and the secondary. If it is desired to transfer the current or voltage information measured in the secondary to stabilize the output characteristics to the primary side, it must be conveyed through an opto-isolator or a corresponding component which realizes galvanic isolation. The price and limited reliability and life of the opto-isolator are disadvantageous factors from the manufacturing standpoint.
The development of switched-mode power supplies has been towards concentrating the control and limiting functions in the primary part only. The idea is based on the fact that the output characteristics can be determined on the basis of certain parameters of the primary part. It is known to add to a transformer like the one described a third, or additional, winding to generate an image of the voltage waveform induced in the secondary winding during one cycle. It is also known to measure the primary current by connecting a small current measuring resistor in series with the switching element and measuring the voltage loss across said resistor. It is beneficial to concentrate the control and limiting functions in the primary because most of the necessary measuring and adjusting connections can be integrated in one IC which also advantageously contains means to produce the switching element ON and OFF signals. The switching element can also be integrated in the same circuit. The disadvantages of opto-isolators are thus avoided.
FIG. 2 illustrates a known method to limit the output current I.sub.OUT in a flyback-type switched-mode power supply. Transformer T1 includes a primary winding 11, a secondary winding 12 and an additional winding 13. The primary side includes a MOSFET transistor Q1 used for switching the primary current, a control circuit F1 to control the gate voltage of said transistor, a current measuring resistor R.sub.s, a differential amplifier A1, a constant current supply I.sub.c, a so-called "external" capacitor C.sub.ext, a current path to discharge said capacitor including a resistor R.sub.c and a switch S1, and a so-called additional circuit 10. The word "external" is used throughout this text simply to qualify the capacitor C.sub.ext and it does not necessarily mean that said capacitor is physically located apart from the rest of the circuit. The additional circuit 10 connected to the additional winding 13 is a detector the task of which is to detect the demagnetization of transformer T1, or the moment at which the energy stored in the magnetic field during one ON cycle of transistor Q1 is completely transferred to the secondary of the power supply. The secondary of the power supply according to FIG. 2 comprises a diode D1, a relatively high-capacity capacitor C1 to stabilize output voltage variation in one cycle, and a relatively large shunt resistor R.sub.o the purpose of which is to serve as a minimum load and to provide a discharge path for the charge stored in capacitor C1 upon switch-off.
The circuit depicted in FIG. 2 operates as follows: At the beginning of a cycle, the control circuit F1 switches the MOSFET transistor Q1 ON, i.e. into conductive state. An increasing primary current I.sub.p starts to flow through the primary winding 11, MOSFET transistor Q1 and the current measuring resistor R.sub.s. Affected by the inductance of winding 11, the primary current I.sub.p increases linearly. The polarities of windings 11 and 12 and diode D1 are such that the magnetic field produced by the primary current I.sub.p tries to induce in the secondary winding 12 a voltage in relation to which the diode D1 is reverse-biased. The diode prevents the flow of current in the secondary circuit, whereby energy is stored in the strengthening magnetic field. When the MOSFET transistor is switched OFF, the sign of the time derivative of the magnetic field is reversed and a current is induced in the secondary winding in relation to which the diode D1 is forward-biased. Part of the secondary current I.sub.s produced is taken to the load as output current I.sub.OUT and part of it charges the capacitor C1, which has maintained the output current I.sub.OUT by partly discharging while the secondary current I.sub.s was not flowing. When the energy in the magnetic field has been completely discharged, the secondary current I.sub.s stops flowing and the additional circuit 10, which measures the small current I.sub.a induced in the additional winding, detects the situation and informs the control circuit F1. The signal provided by the additional circuit affects the pulse ratio: the less time was used to discharge the magnetic energy, the longer the ON time in proportion to the OFF time that is needed in the next cycle and vice versa.
During the ON cycle of the MOSFET transistor Q1 the primary current I.sub.p flows through a current measuring resistor R.sub.s, as described above. A differential amplifier A1 compares the existing voltage loss of resistor R.sub.s to a reference voltage U.sub.ext between the terminals of the external capacitor C.sub.ext. If the existing voltage loss of resistor R.sub.s becomes greater than voltage U.sub.ext, the differential amplifier A1 informs control circuit Fi that the primary current has reached its peak value, whereupon the MOSFET transistor Q1 is immediately switched OFF.
In addition to controlling the gate voltage of the MOSFET transistor Q1 the control circuit F1 controls the position of switch S1. Switch S1 should always be ON, i.e. conductive, when diode D1 is conductive, and correspondingly OFF when diode D1 is non-conductive. This is to keep the voltage U.sub.ext between the terminals of capacitor C.sub.ext correct by providing a current path, which comprises a switch S1 and a series resistor R.sub.c, discharging the capacitor when switch S1 is ON. The control circuit F1 sets switch S1 ON when switching the MOSFET transistor Q1 OFF, and OFF when the additional circuit 10 informs that the transformer T1 has been demagnetized.
By using the demagnetizing information to control the pulse ratio and by preventing the primary current I.sub.p from increasing too much the primary part of the switched-mode power supply shown in FIG. 2 controls the output current I.sub.OUT of the device. When the output voltage U.sub.OUT increases, e.g. when a battery connected to the charger is being charged, the demagnetization time of transformer TI and the conduction time of diode D1 become shorter. The power supply meets the growing demand for output current by increasing the pulse ratio on the basis that also the ON time of switch S1 is shortened, whereupon the voltage U.sub.ext between the terminals of capacitor C.sub.ext is increased. In the opposite case, slow demagnetization or very high primary current mean excessive output current, with switch S1 in conductive state for a long time during the cycle, which decreases the voltage U.sub.ext between the terminals of capacitor C.sub.ext, which in turn decreases the pulse ratio. As regards e.g. a battery charger like the one described above, the disadvantage of this known arrangement is that it includes no output voltage limiter.
FIG. 3a shows a known circuit used for estimating the output voltage of a flyback type switched-mode power supply. The circuit is intended to be used as part of the primary of the power supply, but for reasons of clarity the rest of the primary, with the exception of primary winding 11, are not shown. The voltage regulating circuit (as it will be called hereinafter) according to FIG. 3a comprises a rectifying diode D2, series resistor R2, resistors R3 and R4 for voltage division, capacitor C2 for stabilizing voltage variations at point A during one cycle, and as a load, a constant resistance R.sub.a, which could be replaced by a constant current load the current I.sub.a of which would be advantageously about 10 mA. The voltage regulating circuit also includes a voltage comparison stage which comprises a differential amplifier A2.
The voltage regulating circuit according to FIG. 3a is a kind of a mirror image of the secondary of the switched-mode power supply and indeed it is designed to produce a so-called image voltage U.sub.a at point A in the same way that the secondary produces an output voltage U.sub.OUT at the output terminals of the switched-mode power supply. The image voltage U.sub.a is measured via a voltage divider comprising resistors R3 and R4 and compared to an accurate reference voltage U.sub.ref by means of a differential amplifier A2. The output of the differential amplifier A2 is used to control the primary current switch (not shown) through a pulse width modulator and a control circuit (not shown). The component values are selected such that the power consumption in the voltage regulating circuit is as small as possible. Unfortunately this goal conflicts with the objective that the image voltage U.sub.a should be a perfect image of the controlled output voltage U.sub.OUT.
The fundamental flaw in the operation of the circuit is related to the magnetic properties of transformer T1. In a typical switched-mode power supply transformer that includes an additional winding, such as transformer T1 in FIG. 3a, there has to be an insulating layer between the primary 11 and secondary 12 windings. However, the additional winding 13 can be wound directly over or under the primary winding 11 or even interleaved with it. Therefore, the coupling coefficient between the primary winding 11 and the additional winding 13 is nearer to 1 than the coupling coefficient between the primary winding 11 and the secondary winding 12. In the case of the transformer shown in FIG. 3a, typical coupling coefficients can be about 0.99 between the primary and additional windings and about 0.98 between the primary and secondary windings. The difference of the coupling coefficients means that when the current through the primary winding 11 is shut off, the energy stored by its leak inductance cannot be transferred to the secondary winding 12, but part of it can be transferred to the additional winding 13, which results in a voltage peak across the additional winding. The height of the peak depends on the mount of energy stored in the leak inductance of the primary winding. If the output power, or output current, of the switched-mode power supply is high, a lot of energy is transferred in one cycle and, correspondingly, more energy is lost in the leak inductances than if the output power were low. A great amount of energy stored in the leak inductance means a high voltage peak across the additional winding 13. Capacitor C2 partly rounds off the effect of the voltage peak, but in any case it leads to a nearly linear dependence between the image voltage U.sub.a and the output voltage, as seen in FIG. 4.
A solution to the problem described above would be to add an insulating layer between the primary 11 and additional 13 windings in transformer T1, whereby the coupling coefficient between them would be equal to that between the primary and secondary windings. A sandwich type transformer could also be used. However, the transformer is already the most expensive single component in a switched-mode power supply, so it is not preferable to make its construction more complex.
Another solution is to eliminate the voltage peak using any arrangement known to one skilled in the art. One such arrangement is shown in FIG. 3b where a low-pass type coupling consisting of a resistor R5 and a capacitor C3 is added to the arrangement shown in FIG. 3a. A rectifying diode D3 and a capacitor C4 are also added to the circuit. Resistors R3 and R4 comprise a voltage dividing coupling connected in series with diode D3, used for taking a voltage signal to the differential amplifier A2, said voltage signal being proportional to voltage U.sub.b at point B, i.e. to the rectification result of the low-pass filtered image voltage. Voltage U.sub.b is a better representation of the switched-mode power supply output voltage U.sub.OUT as a function of the output current I.sub.OUT, as can be seen from the curves in FIG. 4, but it still depends, and non-linearly depends, on the output era-rent I.sub.OUT.
The curves in FIG. 4 are the result of a laboratory measurement wherein the output voltage U.sub.OUT of the switched-mode power supply was kept constant and image voltage alternatives U.sub.a and U.sub.b were studied as the function of the output current I.sub.OUT. In a real switched-mode power supply using a voltage regulating circuit according to FIG. 3a or 3b, a differential amplifier A2 compares voltage U.sub.a or U.sub.b to an accurate and constant reference voltage U.sub.ref, so it is implied that the image voltage U.sub.a or U.sub.b, whichever is used, reflects the output voltage U.sub.OUT realistically, being directly proportional to it and wholly independent of the output current I.sub.OUT. Since this is not the case, the output voltage U.sub.OUT of the switched-mode power supply becomes too high with a small output current I.sub.OUT.
Several other ways to improve the output characteristics of a switched-mode power supply are known. A known arrangement includes a sampling circuit, which does not measure the shape of the whole voltage pulse induced in the additional winding, but takes a narrow sample from it, advantageously near the trailing edge of the pulse. The effect of the voltage peak mentioned above is at its smallest near the trailing edge of the pulse. The voltage sample is used to generate an image voltage, which is used in the same manner as image voltages U.sub.a and U.sub.b described above. The sampling circuit naturally adds to the complexity, manufacturing costs and power consumption of the arrangement. It is further known a solution in which a current integrator used for measuring the output voltage is added to the current limiting arrangement according to FIG. 2. The solutions described have not been able to correct the non-linear dependence between the output voltage and output current in a switched-mode power supply, which typically manifests itself in a sharp increase in the output voltage when the output current is small.
It is also known a method called burst mode control, where an image voltage (above, U.sub.a and U.sub.b) is measured and compared to a reference value. If the measured value is greater than the reference value indicating the allowed maximum, a Schmitt trigger circuit grounds the gate of the MOSFET transistor functioning as a primary current switch, i.e. switches the transistor OFF for a predetermined period of time the duration of which is determined by the component values. When the forced grounding of the transistor gate ceases, the cycle starts over again. The problem with this arrangement is that when the forced grounding of the switching transistor gate is ended, the switched-mode power supply immediately starts operating at a high power, pumping a great amount of magnetic energy into the transformer and causing sharp voltage peaks in the voltage across the additional winding. This has the same effect as the fact that the image voltages U.sub.a and U.sub.b mentioned above depend on the output current of the switched-mode power supply: the open circuit voltage, or the output voltage of the device increases when the output current is small.